System, method and apparatus for phase hits and microphonics cancellation

ABSTRACT

A system and method for system, method and apparatus for phase hits and microphonics cancellation. In addition to a first RF synthesizer source, a device also includes a second stable reference signal source that operates at a lower frequency as compared to the RF synthesizer source. The second stable reference signal source is selected with good phase noise characteristics and can be used to correct phase error events.

This application is a continuation of U.S. patent application Ser. No.14/614,792 filed Feb. 5, 2015, which claims priority to U.S. provisionalpatent application 62/087,586, filed on Dec. 4, 2014, now expired. Theseapplications are hereby incorporated herein by reference in theirentirety.

BACKGROUND

Field

The present disclosure relates generally to microwave backhaularchitecture, including a system, method and apparatus for phase hitsand microphonics cancellation.

Introduction

Conventional microwave backhaul architectures are generally implementedas either a split outdoor unit (split ODU) configuration or an alloutdoor unit (all ODU) configuration. Conventional split ODUconfigurations are generally comprised of both an indoor unit (IDU) andan outdoor unit (ODU), where the IDU and the ODU are connected over asingle channel coaxial interconnect. The IDU in a conventional split ODUconfiguration typically includes a modem, a digital-to-analog converterand a baseband-to-intermediate frequency converter.

Mobile backhaul providers are experiencing a growing demand forincreased capacity as well as a shift from voice services to dataservices. These factors are driving mobile backhaul networks towardshigh capacity IP/Ethernet connections. Additionally, the transition to4G and LTE networks is also driving the need for higher capacity, andmoving more packet traffic onto mobile backhaul networks.

BRIEF DESCRIPTION OF THE DRAWINGS

In order to describe the manner in which the above-recited and otheradvantages and features can be obtained, a more particular descriptionwill be rendered by reference to specific embodiments thereof which areillustrated in the appended drawings. Understanding that these drawingsdepict only typical embodiments and are not therefore to be consideredlimiting of its scope, the disclosure describes and explains withadditional specificity and detail through the use of the accompanyingdrawings in which:

FIG. 1 illustrates a block diagram of a microwave backhaul systemaccording to an exemplary embodiment.

FIG. 2A illustrates a high-level block diagram of an IDU forimplementation within a microwave backhaul system according to anexemplary embodiment.

FIG. 2B illustrates a high-level block diagram of an ODU forimplementation within a microwave backhaul system according to anexemplary embodiment.

FIG. 3 illustrates an example of a mechanism for correcting phase errorevents.

FIG. 4 illustrates another example of a mechanism for correcting phaseerror events.

FIG. 5 illustrates an example embodiment of a mechanism for correctingphase error events in a receiver.

FIG. 6 illustrates an example embodiment of a mechanism for correctingphase error events in a transmitter.

FIG. 7 illustrates a flowchart of an example process.

DETAILED DESCRIPTION

Various embodiments are discussed in detail below. While specificimplementations are discussed, it should be understood that this is donefor illustration purposes only. A person skilled in the relevant artwill recognize that other components and configurations may be usedwithout parting from the spirit and scope of the present disclosure.

It is recognized that the microwave backhaul world is growing due toincreased bandwidth demands. FIG. 1 illustrates a block diagram of anexample microwave backhaul system 100 that includes IDU 102 and ODU 104.Microwave, as used throughout this disclosure, refers to bothterrestrial point-to-point (PtP) radio communications, as well aspoint-to-multipoint communications, and can include both wired and/orwireless communications.

Microwave backhaul system 100 initiates communication by accessing aninformation source, which can comprise, for example, audio data 106,video data 108, or any other data capable of being transmitted over anInternet Protocol (IP)/Ethernet connection 110. To facilitate thiscommunication, IDU 102 can be coupled to a core network. In particular,IDU 102 can be configured to acquire one or more sequences of digitaldata (e.g., audio data 106, video data 108, data transmitted overIP/Ethernet connection 110, or the like) from the core network. IDU 102can also be configured to support several additional services, such asEthernet, time-division multiplexing (TDM), and control data that isaggregated over a radio link.

IDU 102 can be implemented at a location that is substantially removedfrom ODU 104, such as at a location at ground level. For example, IDU102 can be positioned inside of a home or an office building, or otherstructure. Conversely, ODU 104 can be implemented at a substantiallyelevated location, such as on top of a pole, on top of an antenna tower,on top of a building, or other mounted location. In some embodiments,IDU 102 and ODU 104 can be separated by a significant distance (e.g., upto approximately 300 meters). In general, IDU 102 includes a modernassembly, while ODU 104 includes at least some RF functionalities aswell as corresponding digital capabilities.

IDU 102 and ODU 104 can be connected via communication pathway 112,which can be configured such that data can be transmitted between IDU102 and ODU 104. In various examples, communication pathway 112 cancomprise a twisted pair Ethernet cable, a fiber optic cable, a coaxialcable, an intermediate frequency (IF) cable, or any other cable suitablefor IDU-ODU communication. Therefore, depending on a chosencommunication medium, communication pathway 112 can facilitatetransmission of an analog signal or a digital signal between IDU 102 andODU 104. In some embodiments, communication pathway 112 can be awireless communication channel.

Antenna 116 can be coupled to ODU 104, and can be positioned close toODU 104. Therefore, microwave backhaul system 100 can be implementedsuch that data can be transmitted from IDU 102, across communicationpathway 112, to ODU 104, and subsequently to antenna 116 wherecommunication over a wireless link can then be initiated. Also,microwave backhaul system 100 can be implemented such that data receivedby antenna 116 can be transmitted from ODU 104 over communicationpathway 112 to IDU 102.

In one embodiment, ODU 104 can correct errors associated with a signalreceived over a wireless link via antenna 116. Microwave backhaul system100 can also be configured to support adaptive coding and modulation(ACM), which provides high reliability of microwave backhaul system 100even in extreme weather, such as wind, rain, hail, or other interferingenvironmental conditions. For example, ACM can adapt coding andmodulation rates to changing environmental conditions to therebyincrease throughput over a link and make efficient use of the existingspectrum. Thus, ACM enables the ODU to hitlessly manage the transitionswhen adjusting the number of transmission/receipt channels based on thechanges in the communication channel

FIGS. 2A and 2B illustrate high-level block diagrams of an example IDUand ODU, respectively, for use within a microwave backhaul systemaccording to an exemplary embodiment. IDU 202 and ODU 204 are coupledtogether via communication pathway 212. IDU 202 can represent anexemplary embodiment of IDU 102 of FIG. 1, and ODU 204 can represent anexemplary embodiment of ODU 104 of FIG. 1.

IDU 202 includes a power supply unit (PSU) 206, a CPU 208, a modemassembly 210, a digital-to-analog converter/analog-to-digital converter(DAC/ADC) block 216, a modulation block 218, and an intermediatefrequency (IF) module 220. In some embodiments, IDU 202 can also includean N-Plexer 222. CPU 208 is configured to carry out instructions toperform arithmetical, logical, and/or input/output (I/O) operations ofone or more of the aforementioned elements contained within IDU 202. Inan embodiment, CPU 208 can control operation of modulation block 218 andN-Plexer 222.

Modem assembly 210 is configured to perform modulation and demodulationof data that is to be transmitted between IDU 202 and ODU 204. In someembodiments, modem assembly 210 can function substantially similar to abaseband modem. Further, modem assembly 210 can be configured to cancelout noise associated with IDU 202 or communication pathway 212. DAC/ADCblock 216 can be configured to transmit and/or receive data from modemassembly 210. DAC/ADC block 216 is also configured to performdigital-to-analog and/or analog-to-digital conversions of data such thatthe data is suitable for transmission over communication pathway 212.

Modulation block 218 can also be configured to perform variousmodulation and/or demodulation techniques. In an embodiment, modulationblock 218 can be configured to perform amplitude-shift keying. Forexample, modulation block 218 can be configured to performamplitude-shift keying by utilizing a finite number of amplitudes, whereeach amplitude is assigned a unique pattern of binary digits. Eachpattern can then be configured to form the specific symbol that isrepresented by the particular amplitude. Additionally, when modulationblock 218 is configured to perform demodulation, modulation block 218determines the amplitude of the received signal and maps it back to thesymbol it represents, thus recovering the original data.

IF module 220 can be configured to transmit and/or receive data fromDAC/ADC block 216. IF module 220 is also configured to perform afrequency conversion of the received data such that data is suitable fortransmission over communication pathway 212. For example, IF module 220can be configured to convert data from a baseband frequency to an IF.

N-Plexer 222 can be configured to permit N-directional communicationover communication pathway 212. In particular, N-Plexer 222 isconfigured to isolate IDU 202 from ODU 204, while permitting them toshare a common antenna. N-Plexer 222 is also configured to receive acontrol signal (e.g. a Telemetry ASK signal) output from modulationblock 218, and to receive an IF signal output from IF module 220.Additionally, N-Plexer 222 can be configured to convert and/or combineeach of these inputs to form data. N-Plexer 222 is also configured totransmit and/or receive data, over communication pathway 212, betweenIDU 202 and ODU 204. In an embodiment, N-Plexer 222 can functionsubstantially as an analog duplexer (multiplexer/demultiplexer). In oneembodiment, communication pathway 212 can be embodied as one or more IFcables that can facilitate quadruple channel communication with one ormore IDUs.

In an exemplary embodiment, DAC/ADC block 216, modulation block 218, IFmodule 220 and N-Plexer 222 can be replaced by Digital N-Plexer 226. Inparticular, Digital N-Plexer 226 can be configured tomultiplex/demultiplex the required signal in the digital domain, ratherthan in the analog domain. Subsequently, Digital N-Plexer 226 can allowcommunication pathway 212 to be implemented as either a digital pathwayor an analog pathway. Using Digital N-Plexer 226 allows for a simplerimplementation of IDU 202. For example, when implementing IDU 202 havingDigital N-Plexer 226, no analog functionality would be required, andinstead only a single digital chip substrate would be needed. As aresult, the cost of implementing IDU 202 can be decreased. Additionally,using a Digital N-Plexer 226 can provide an improved yield, shorterproduction testing, lower assembly cost, lower peripheral componentcount, and can support greater distances between IDU 202 and ODU 204, toprovide some examples.

As illustrated in FIG. 2B, ODU 204 can also include an N-Plexer 228,which can be implemented in several different manners. For example,N-Plexer 228 can be an analog N-Plexer, a digital N-Plexer, or a splitfunction N-Plexer (e.g., where N-Plexer 228 is partially analog andpartially digital). When N-Plexer 228 represents a digital N-Plexer,N-Plexer 228 can function in a substantially similar manner as DigitalN-Plexer 226. In particular, N-Plexer 228 can be configured tomultiplex/demultiplex signals in the digital domain. N-Plexer 228 alsoallows for a simpler implementation of ODU 204 because no analogfunctionality would be required, and instead only a single digital chipsubstrate would need to be implemented within ODU 204. Therefore, thecost of implementing ODU 204 can also be decreased. Similar to DigitalN-Plexer 226, implementing N-Plexer 228 within ODU 204 can provide animproved yield, shorter production testing, lower assembly cost, lowerperipheral component count, and can support greater distances betweenIDU 202 and ODU 204, to provide some examples.

In an embodiment, IDU 202 and ODU 204 can be configured to perform anN-Plexer elimination technique. In particular, the functionalitydirected towards filtering RX signals, after being received overcommunication pathway 212, and TX signals, before being transmitted overcommunication pathway 212, can be removed from N-Plexers 226 and 228.Instead, this functionality can be implemented within the digital chipsubstrate (e.g., integrated circuit) that comprises IDU 202 and thedigital chip substrate (e.g., integrated circuit) that comprises ODU204. IDU 202 and ODU 204 can then filter the required signals throughany combination of an analog filtering process, a signal samplingprocess and/or a digital filtering process.

ODU 204 can also include CPU 230, ADC/DAC blocks 232 and 236, digitalsignal processor (DSP) 248, and RF module 234. CPU 230 can be configuredto function in a substantially similar manner as CPU 208. In particular,CPU 230 can be configured to carry out instructions to performarithmetical, logical, and/or I/O operations of one or more of theelements contained within ODU 204. In an embodiment, CPU 208 can controloperation of N-Plexer 228. ADC/DAC block 232 can be configured totransmit and/or receive data from N-Plexer 228. ADC/DAC blocks 232 and236 are also configured to perform analog-to-digital and/ordigital-to-analog conversions of data such that data can be properlytransmitted and/or received over communication pathway 212. Further, DSP248 can be configured to perform mathematical manipulation techniques ondata, such that data may be modified or improved according to a desiredprocessing method. For example, DSP 248 can be configured to measure,filter, or compress data prior to being output to ADC/DAC block 236,such that error detection and/or error correction can be performed onthe data. In an embodiment, after the data is received, overcommunication pathway 212, at ODU 204, the data traverses throughN-Plexer 228, to ADC/DAC block 232, to DSP 248, to ADC/DAC block 236, toRF module 234 and to antenna 244 before being transmitted acrosswireless link 246. Similarly, after data is received over wireless link246, at ODU 204, data traverses from antenna 244 to RF module 234, toADC/DAC block 236, to DSP 248, to ADC/DAC block 232, and to N-Plexer 228before being transmitted over communication pathway 212. As will bedescribed in greater detail below, DSP 248 can also be configured toimplement a phase hits and microphonics cancellation mechanism.

RF module 234 can be configured to transmit and/or receive data fromADC/DAC block 236. RF module 234 can also be configured to perform afrequency conversion of data such that data can be properly receivedover communication pathway 212. For example, when data is received at RFmodule 234, data can have a frequency residing in the IF range.Therefore, RF module 234 can up-convert data from an IF to a RF suchthat data can then be transmitted over wireless link 246. RF module 234can also be configured to down-convert a signal received over thewireless link from a RF to an IF such that the received signal can betransmitted over communication pathway 212 to IDU 202.

As noted, there is a continuing need for higher capacity systems usinghigher levels of modulation. For example, work is ongoing to extend thecapacity afforded by 4096 quadrature amplitude modulation (QAM) systemsto higher capacity systems using 16,384 QAM. Additional capacity growthcan also be enabled using more complex systems that leverage crosspolarization and 4×4 spatial multiple-input-multiple-output (MIMO)systems.

In the present disclosure, it is recognized that phase hits andmicrophonics is a major impediment to the development of high modulationand MIMO systems. These systems are sensitive to phase hits under 1 kHz.When microwave local oscillator (LO) frequencies are in the range of 6GHz to 43 GHz, phase hits and microphonics makes it nearly impossible toreach zero bit error rate (BER) performance in high modulation and MIMOsystems.

The present disclosure presents a mechanism for phase hits andmicrophonics cancellation that can be used to enable high modulation andMIMO systems. FIG. 3 illustrates an example of a mechanism forcorrecting phase error events. As illustrated, the microwave deviceincludes an analog transmitter or receiver section 310 that isconfigured to operate based on a LO signal generated by RF synthesizer330. The signal generated by RF synthesizer 330 can be characterized asS_(LO)(t)=A_(LO) cos(ω_(LO)·t+φ_(LO)(t)). As would be appreciated, thefrequency of the LO signal (e.g., 8 GHz) can vary depending on theimplementation and application.

As further illustrated in FIG. 3, the microwave device also includes asecond signal reference 350. In the present disclosure, it is recognizedthat signal reference 350 can be established at a significantly lowerfrequency (e.g., 800 MHz) as compared to RF synthesizer 330 and cantherefore be implemented as a stable reference with very good phasenoise characteristics. In one example, signal reference 350 can beembodied as a voltage controlled oscillator (VCO) that has a very high Qand is not necessarily frequency locked. In other examples, signalreference 350 can be free running from any clock source such as atemperature-controlled crystal oscillator (TCXO), ceramic resonatoroscillator (CRO), surface acoustic wave (SAW) resonator, or any otherstable reference source that does not suffer from phase hits. In yetanother example, a VCO can be placed physically in a so called“protected” environment where both temperature and voltage are wellmonitored. Here, it should be recognized that optimal voltage andtemperature conditions can be achieved in the IDU.

The signal generated by signal reference 350 can be characterized asS_(REF)(t)=A_(REF) cos(ω_(REF)·t+φ_(REF)(t)). As would be appreciated,the particular frequency of operation of signal reference 350 can varyand can be selected to enable phase cancellation relative to a dividedRF synthesizer 330 signal.

In the example of FIG. 3, the signal output by RF synthesizer 330 isdivided by a factor N by divider 340. In one example, divider 340 candivide an 8 GHz signal output by RF synthesizer 330 by a factor of 10 toproduce an 800 MHz signal. In one example, signal reference 350 can beselected to generate an 800 MHz signal.

The signal output by divider 340 is provided as an input toanalog-to-digital converter (ADC) 322 in a digital core of the microwavedevice. Similarly, the signal output by signal reference 350 is providedas an input to ADC 324 in the digital core of the microwave device.Phase detector 323 is configured to detect the phase in the digitalsignal representative of the divided LO signal generated by RFsynthesizer 330. Similarly, phase detector 325 is configured to detectthe phase in the digital signal representative of the signal generatedby signal reference 350.

The phase of the divided LO signal as determined by phase detector 323can be subtracted from the phase of the reference signal as determinedby phase detector 324 using subtraction module 326. This comparison bysubtraction module 326 of the phases of the divided LO signal and thereference signal enables a determination of a phase error of RFsynthesizer 330 in comparison to signal reference 350. Subtractionmodule 326 can then generate a phase correction signal, which can berepresented as S_(phase) _(_) _(corr)[n]≅N·φ_(REF)[n]−φ_(LO)[n−τ]. Asillustrated, a phase correction signal output by subtraction module 326is applied to phase rotation module 327, which performs a rotation onthe data path to correct the phase error events. Here, it is recognizedin the present disclosure that as long as signal reference 350 is astable signal source, the device can achieve phase hits and microphonicsimmunity.

The example of FIG. 3 provides a phase hits and microphonicscancellation mechanism using differential sampling of a LO signal and areference signal. In another example, a phase hits and microphonicscancellation mechanism is enabled by mixing of two products, wherein afirst product is sourced from an RF synthesizer and a second product issourced from a signal reference. FIG. 4 illustrates this other examplemechanism for correcting phase error events.

As illustrated, the microwave device includes an analog transmitter orreceiver section 410 that is configured to operate based on a LO signalgenerated by RF synthesizer 430. The signal generated by RF synthesizer430 can be characterized as S_(LO)(t)=A_(LO) cos(ω_(LO)·t+φ_(LO)(t)). Aswould be appreciated again, the frequency of the LO signal (e.g., 8 GHz)can vary depending on the implementation and application.

As further illustrated in FIG. 4, the microwave device also includes asecond signal reference 450. Again, it is recognized in the presentdisclosure that signal reference 450 can be established at asignificantly lower frequency (e.g., 800 MHz) as compared to RFsynthesizer 430 and can therefore be implemented as a stable referencewith very good phase noise similar to RF synthesizer 330. The signalgenerated by signal reference 450 can be characterized asS_(REF)(t)=A_(REF) cos(ω_(REF)·t+φ_(REF)(t)). As would be appreciated,the particular frequency of operation of signal reference 450 can varyand can be selected to enable phase cancellation relative to a dividedRF synthesizer 430 signal.

In the example of FIG. 4, a signal sourced from RF synthesizer 430 ismixed with a reference signal sourced from signal reference 450 by mixer460. The signal output by mixer 460 is provided as an input to ADC 422in a digital core of the microwave device. Phase detector 423 is thenconfigured to detect the phase error of RF synthesizer 430 relative tosignal reference 450. Phase detector 423 can then generate a phasecorrection signal, which can be represented as S_(phase) _(_)_(corr)[n]≅φ_(REF)[n]−φ_(LO)[n−τ]. The phase correction signal output byphase detector 423 is then applied to phase rotation module 427, whichperforms a rotation on the data path to correct the phase error events.Here again, it is recognized in the present disclosure that as long assignal reference 350 is a stable signal source, the device can achievephase hits and microphonics immunity.

FIG. 5 illustrates an example embodiment of a mechanism for correctingphase error events in a receiver using the mixing of two products,wherein a first product is sourced from an RF synthesizer (RF LO) and asecond product is sourced from a signal reference (VCO2) established ata significantly lower frequency as compared the RF synthesizer. Asillustrated, RF LO 530 generates an LO signal that is used for adown-conversion mixer in a conventional receive chain of a microwavedevice.

In the illustrated embodiment, the LO signal generated by RF LO 530 isdivided by N then mixed with the signal generated by VCO2 550. Afterboth division by N and being further down-converted with the signalgenerated by VCO2 550, the signal frequency is sufficiently low suchthat its phase can be extracted. Next, the extracted phase passesthrough a band pass filter (BPF) having a passband [F₁÷F₂].

In one embodiment, the high pass filter (HPF) part of the BPF filtersout the carrier frequency of the phase. Since the frequency of VCO2 550is not locked to an arbitrary reference the frequency after the phaseextraction is meaningless. In one embodiment, the value of F₁ can be setaccording to the following objectives. First, the F₁ minimal value canbe sufficiently high to track down the frequency drift of VCO2 550. Thefrequency drift can be caused mainly by temperature and aging andtherefore a few Hz should be sufficient for tracking. Second, the F₁maximal value should be sufficiently low such that the modem phasetracking circuits can cancel the phase in the range [0÷F₁]. For example,assuming that pilots spacing=41 and F_(sym)=6 Msym/s the Pilot-SymbolAssisted Modulation (PSAM) can cancel at least 20 kHz.

In one embodiment, the low-pass filter (LPF) part of the BPF limits thecorrected phase to low frequencies only since correction in highfrequencies is not possible due to very low values of the phase noiseand sensitivity to delay mismatches. The corner frequency F₂ can be setaccording to the following objectives. First, the F₂ minimal value canbe sufficiently high to filter out the phase hits. It is recognized thatthe required bandwidth can be 1÷2 MHz but the precise value can dependon the phase hit model (if any). Second, the F₂ maximal value can besufficiently low such that cancellation of phase noise is valid forf<F₂. As would be appreciated, the precise value can depend on specificsystem parameters such as the phase noise of RF LO 530, VCO2 550 and thedelay resolution.

After the BPF the phase is subtracted from the signal phase. Since thesignal phase includes the RF synthesizer phase, subtracting the phaseshould cancel out the RF synthesizer phase. Mathematically, the phasecancellation looks as follows in equation (1):

${\phi_{out}(t)} = {{K \cdot {\phi_{RFLO}(t)}} - {{KN} \cdot h_{{BPF}{(t)}} \cdot \left( {\frac{\phi_{RFLO}(t)}{N} - {\phi_{{VCO}\; 2}(t)}} \right)}}$

Rewriting equation (1) in the frequency domain leads to the following inequation (2):

${\phi_{out}(f)} = {{{K \cdot {\phi_{RFLO}(f)}} - {{{KN} \cdot {H_{BPF}(f)}}\left( {\frac{\phi_{RFLO}(f)}{N} - {\phi_{{VCO}\; 2}(f)}} \right)}} = {{{K \cdot {\phi_{RFLO}(f)}}\left( {1 - {H_{BPF}(f)}} \right)} + {{KN} \cdot {H_{BPF}(f)} \cdot {\phi_{{VCO}\; 2}(f)}}}}$

Summarizing equation (2) it can be seen that the original phase noise(without correction) K·ϕ_(RFLO)(f) is filtered by 1−H_(BPF)(f) whichdecreases its variance substantially. VCO2 sets a floor on the phasenoise variance which emphasizes the importance of choosing VCO2 with alow phase noise. The tradeoff in choosing H_(BPF)(f) can also be deducedfrom equation (2). Choosing a wide bandwidth for H_(BPF)(f) providesbetter filtering for ϕ_(RFLO)(f) but increases the contribution ofϕ_(VCO2)(f).

The optional delay in FIG. 5 is used for compensating the processingtime of the phase correction branch which can be longer then the mainsignal chain due to the narrow bandwidth of H_(BPF)(f). Addingadditional delay of τ to equation (2) produces the following in equation(3):

${\phi_{out}(f)} = {{{{K \cdot {\phi_{RFLO}(f)}}e^{j\; 2\pi\;{f \cdot \tau}}} - {{{KN} \cdot {H_{BPF}(f)}}\left( {\frac{\phi_{RPLO}(f)}{N} - {\phi_{{VCO}\; 2}(f)}} \right)}} = {{{K \cdot {\phi_{RFLO}(f)}}\left( {e^{j\; 2\pi\;{f \cdot \tau}} - {H_{BPF}(f)}} \right)} + {{KN} \cdot {H_{BPF}(f)} \cdot {\phi_{{VCO}\; 2}(f)}}}}$

Here, the delay τ can be optimized to produce minimal variance forϕ_(OUT)(f).

FIG. 6 illustrates an example embodiment of a mechanism for correctingphase error events in a transmitter using the mixing of two products,wherein a first product is sourced from an RF synthesizer (RF LO) and asecond product is sourced from a signal reference (VCO2) established ata significantly lower frequency as compared the RF synthesizer. Asillustrated, the schematics for the transmitter is similar to thereceiver embodiment illustrated in FIG. 5. One difference is that thetransmitter cannot compensate for the phase correction branch delaysince the correction is always applied with delay relative to theoriginal phase noise. Mathematically, this changes equation (2) into thefollowing in equation (4), where T_(D) is the delay of the feedbackphase correction branch:

${\phi_{out}(f)} = {{{K \cdot {\phi_{RFLO}(f)}} - {{KN} \cdot {H_{BPF}(f)} \cdot {e^{j\; 2\pi\;{f \cdot T_{D}}}\left( {\frac{\phi_{RFLO}(f)}{N} - {\phi_{{VCO}\; 2}(f)}} \right)}}} = {{{K \cdot {\phi_{RFLO}(f)}}\left( {1 - {e^{j\; 2\pi\;{f \cdot T_{D}}} \cdot {H_{BPF}(f)}}} \right)} + {{KN} \cdot e^{{j2}\;\pi\;{f \cdot T_{D}}} \cdot {H_{BPF}(f)} \cdot {\phi_{{VCO}\; 2}(f)}}}}$

As has been described, the addition of an additional frequency sourcehaving very good phase noise can be used to decrease the phase noise ofany LO in a specified bandwidth.

Having described a framework for phase hits and microphonicscancellation in a device, reference is now made to FIG. 7, whichillustrates a flowchart of an example process. As illustrated, theprocess begins at step 702 where a first signal is generated at a firstfrequency. In one example, the first signal can be a LO signal generatedby an RF synthesizer source for use by a microwave device. At step 704,a second signal is generated by a reference source at a second frequencysignificantly lower than the first frequency. In one example, the secondsignal can be generated by a second signal source, which ischaracterized as a stable reference with very good phase noisecharacteristics. At step 706, the first signal is then divided by afactor N.

Next, at step 708, a phase correction signal is generated to address adifference between a phase of the divided first signal and the phase ofthe second signal. In one embodiment, the phase correction signal isbased on differential sampling of the divided first signal and thesecond signal. In another embodiment, the phase correction signal isgenerated based on a mixing of two products, wherein a first product issourced from an RF synthesizer and a second product is sourced from thereference source. Finally, at step 710, the phase correction signal isapplied to a phase rotation module in a data path to correct phase errorevents. In one embodiment, the phase correction signal is applied to aphase rotation module in a transmit data path. In another embodiment,the phase correction signal is applied to a phase rotation module in areceive data path.

In one embodiment, the phase hit/noise cancellation mechanism can beperformed in the baseband modem. In another embodiment, the phasehit/noise cancellation mechanism can be implemented in a standalonesynthesizer without being combined with a baseband modem.

With the phase hit/noise cancellation mechanism, it is expected thatsystems such as 4096 QAM Zero BER systems, 4096 QAM Zero BER CrossPolarization Cancellation (XPIC) systems, 4096 QAM Zero BER 4×4 MIMOsystems, and other higher modulation systems that are sensitive to phasehits and microphonics can be enabled.

Another embodiment of the present disclosure can provide a machineand/or computer readable storage and/or medium, having stored thereon, amachine code and/or a computer program having at least one code sectionexecutable by a machine and/or a computer, thereby causing the machineand/or computer to perform the steps as described herein.

Those of skill in the relevant art would appreciate that the variousillustrative blocks, modules, elements, components, and methodsdescribed herein may be implemented as electronic hardware, computersoftware, or combinations of both. To illustrate this interchangeabilityof hardware and software, various illustrative blocks, modules,elements, components, methods, and algorithms have been described abovegenerally in terms of their functionality. Whether such functionality isimplemented as hardware or software depends upon the particularapplication and design constraints imposed on the overall system. Thoseof skill in the relevant art can implement the described functionalityin varying ways for each particular application. Various components andblocks may be arranged differently (e.g., arranged in a different order,or partitioned in a different way) all without departing from the scopeof the subject technology.

These and other aspects of the present disclosure will become apparentto those skilled in the relevant art by a review of the precedingdetailed disclosure. Although a number of salient features of thepresent disclosure have been described above, the principles in thepresent disclosure are capable of other embodiments and of beingpracticed and carried out in various ways that would be apparent to oneof skill in the relevant art after reading the present disclosure,therefore the above disclosure should not be considered to be exclusiveof these other embodiments. Also, it is to be understood that thephraseology and terminology employed herein are for the purposes ofdescription and should not be regarded as limiting.

What is claimed is:
 1. A device, comprising: a first frequency source ata first frequency; a divider operable to generate an output frequencybased on the first frequency source, wherein the output frequency isnon-zero; a second frequency source at a second frequency, the secondfrequency being lower than the first frequency; and a digital signalprocessor (DSP) operably coupled to the divider and the second frequencysource, wherein a phase correction signal output of the DSP is based ona difference between a detected first phase of the output frequency ofthe divider and a detected second phase of the second frequency, whereinthe phase correction signal output is operable to correct a phase errorexternal to the first frequency source and the second frequency source.2. The device of claim 1, wherein the second frequency is equal to theoutput frequency.
 3. The device of claim 1, wherein the second frequencysource is a ceramic resonator oscillator.
 4. The device of claim 1,wherein the second frequency source is a surface acoustic waveresonator.
 5. The device of claim 1, wherein the second frequency sourceis a free running voltage controlled oscillator.
 6. The device of claim1, wherein the second frequency source is a temperature controlledcrystal oscillator.
 7. A device, comprising: a first frequency source ata first frequency; a divider operably coupled to the first frequencysource to generate an output frequency, wherein the output frequency isnon-zero; a second frequency source at a second frequency, the secondfrequency being lower than the first frequency; a mixer operably coupledto the divider and the second frequency source; and a digital signalprocessor (DSP) operably coupled to the mixer, wherein a phasecorrection signal output of the DSP is based on a filtering of an outputfrom the mixer, wherein the phase correction signal output is operableto correct a phase error external to the first frequency source and thesecond frequency source.
 8. The device of claim 7, wherein the secondfrequency is equal to the output frequency.
 9. The device of claim 7,wherein the second frequency source is a ceramic resonator oscillator.10. The device of claim 7, wherein the second frequency source is asurface acoustic wave resonator.
 11. The device of claim 7, wherein thesecond frequency source is a free running voltage controlled oscillator.12. The device of claim 7, wherein the second frequency source is atemperature controlled crystal oscillator.
 13. A method comprising:generating, using a first frequency source, a first signal at a firstfrequency; generating, using a second frequency source, a second signalat a second frequency lower than the first frequency; dividing the firstsignal at the first frequency to generate a divided first signal,wherein the divided first signal is at a non-zero frequency; generatinga phase correction signal based on a difference between a phase of thedivided first signal and a phase of the second signal; and applying thegenerated phase correction signal to a phase rotation module to correcta phase error, wherein the first frequency source is independent of thegenerated phase correction signal, and wherein the phase error isexternal to the first frequency source and the second frequency source.14. The method of claim 13, wherein the second frequency source is aceramic resonator oscillator.
 15. The method of claim 13, wherein thesecond frequency source is a surface acoustic wave resonator.
 16. Themethod of claim 13, wherein the second frequency source is a freerunning voltage controlled oscillator.
 17. The method of claim 13,wherein the second frequency source is a temperature controlled crystaloscillator.
 18. The method of claim 13, wherein the generating comprisessubtracting a phase of the divided first signal from the phase of thesecond signal.
 19. The method of claim 13, wherein the generatingcomprises mixing the divided first signal with the second signal toproduce a mixed output signal.
 20. The method of claim 19, wherein thegenerating comprises filtering the mixed output signal.